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Texas Instruments Incorporated
Power Management
Get low-noise, low-ripple, high-PSRR
power with the TPS717xx
By Jeff Falin
Senior Applications Engineer
Introduction
While highly efficient switching power supplies are com-
monly used for long battery life in portable end equipment
such as mobile phones and PDAs, the internal circuitry of
some of these devices is sensitive to noise and therefore does
not operate properly when powered from a switching power
supply with output ripple. Audio circuitry, PLLs, RF trans-
ceivers, and DACs are just a few examples of such circuits.
Linear regulators are ideal for powering these circuits.
Figure 1 shows a simplified block diagram of a linear
regulator using a p-channel MOSFET (pFET) as a pass
element. A
OL
is the open-loop gain of the error amplifier,
and g
m
is the pass-element transconductance. The error
amplifier controls the voltage at the gate of the pass element
so that the current through the FET keeps the output
voltage regulated relative to the internal reference voltage.
Assuming that the low-pass filter (LPF) formed by R
LPF
and C
LPF
eliminates nearly all internal-reference noise, the
output voltage should be ripple- and noise-free for frequen-
cies within the bandwidth of the regulator’s control loop.
The concept is easy to understand, but achieving a high
power-supply rejection ratio (PSRR) over a wide bandwidth
with very low quiescent current and in a small package
requires innovative circuits. This article highlights the
TPS717xx single-output linear regulator, which provides
high power-supply rejection (PSR) over a wide bandwidth
with very low quiescent current and in a small package.
Similar, dual-output versions are available with the TPS718xx
and TPS719xx families. This article also provides guidance
on component selection and layout techniques for maxi-
mizing PSR and minimizing the regulator’s self-generated
white noise.
What is the PSRR?
The PSRR is a measure of a circuit’s PSR expressed as a
ratio of output noise to noise at the power-supply input. It
provides a measure of how well a circuit rejects ripple at
various frequencies injected from its input power supply.
In the case of linear regulators, PSRR is a measure of the
regulated output-voltage ripple compared to the input-
voltage ripple over a wide frequency range and is expressed
in decibels (dB). If the pass element in Figure 1 is treated
as a variable resistance, R
DS
, and the error amplifier and
bandgap reference are assumed to have been designed to
Figure 1. Simplified block diagram of a
linear regulator
C
PAR1
C
PAR1
Z
CL
Z
OL
R
DS
Rds
R1
–g
m
C
PAR2
+
R
ESR
A
OL
–
C
OUT
R2
Bandgap
Reference
Voltage
R
LPF
–
V
IN
C
LPF
Quickstart
minimize pass-through of the input-voltage ripple, then
the PSR is simply a voltage divider, expressed as
ZZ
ZZR
||
OL
CL
PSR
=
.
||
+
OL
CL
DS
In this equation, Z
OL
is the output impedance at the
regulator’s output, ignoring the effect of the regulator’s
feedback loop:
Z
=
(
Z
+
R
) || (
RRC
12
+
) ||
,
OL
COUT
ESR
PAR
2
where Z
COUT
and R
ESR
are the output capacitor’s impedance
and equivalent series resistance (ESR), respectively, and
C
PAR2
is the parasitic capacitance of the output components
and PCB. Z
CL
is the impedance looking back into the out-
put of the regulator, including the effect of the regulator’s
feedback loop:
ZRC
gA f
||
||
OL DS PAR
m L
1
β
Z
=
,
CL
×
×
×
where C
PAR1
is the passive-element parasitic capacitance,
f is the ripple frequency, and
β
is the feedback factor,
R
RR
2
1
β=
.
+
17
Analog Applications Journal
3Q 2007
www.ti.com/aaj
High-Performance Analog Products
Power Management
Texas Instruments Incorporated
Figure 2 shows the general shape of a PSRR
curve, where f
P(dom)
is the dominant pole and
f
UG
is the unity-gain bandwidth. If the error
amplifier is compensated to have a single-pole
response, then the Region 1 PSR for amplifier
frequencies below f
UG
can be approximated
by the equation on the left side of the graph.
Designing the regulator with a high-gain,
wide-bandwidth error amplifier can therefore
provide high PSR over a wide range of fre-
quencies. In Region 2, above the control-loop
bandwidth, the regulator is no longer effective
at providing PSR, so the PSRR reduces to a
simple voltage divider as shown on the right
side of the curve. As Z
COUT
decreases relative
to R
DS
, the PSR provided by the passive com-
ponents on the board increases. If C
OUT
has
high R
ESR
, the PSR peaks sooner. In Region 3,
the IC and board parasitic capacitances (C
PA R 1
and C
PAR2
) dominate, resulting in a capacitive
voltage divider, which typically causes the
PSR to decrease again. A larger output capac-
itor with less ESR will typically improve PSRR
in this region, but it can also actually decrease
the PSRR at some frequencies. This occurs because
increasing the output capacitor may lower f
P(dom)
and/or
f
UG
, depending on how the regulator is compensated,
thereby causing the open-loop gain to roll off sooner.
Maximizing PSR
The TPS717xx family of regulators has incorporated both
well-known and patentable circuit techniques to provide
high PSR over a wide frequency range. An example of the
PSRR is shown in Figure 3.
With the simple model previously explained, it can be
shown that the TPS717xx’s dominant pole with C
OUT
= 1 µF
is at approximately 20 to 30 kHz and the unity-gain fre-
quency is near 400 kHz. Since PSR is a function of the open-
loop gain, as the gain varies so will the PSR in Regions 1
Figure 2. PSRR graph
⎡
s
⎤
1
+
⎢
⎢
⎢
⎢
⎢
⎥
⎥
⎥
⎥
⎥
f
⎛
⎜
Z
ZR
⎞
⎟
Pdom
(
)
OL
−
20 log
−
20
log
+
⎛
⎜
s
⎞
⎟
1
+
OL
DS
gA
β
mOL
f
UG
⎣
⎦
Region 1
Region 2
Region 3
If R
ESR
is large
f
P(dom)
f
UG
Frequency, f (log Hz)
and 2 of Figure 2. Figure 3 shows the TPS717xx’s PSRR
varying with load current. As load current increases, R
DS
decreases; therefore Z
CL
decreases, since a MOSFET’s
output impedance is inversely proportional to its drain
current. In many regulators, where f
P(dom)
varies with Z
CL
,
increasing the load current also pushes f
P(dom)
to higher
frequencies, which increases the feedback-loop band-
width. As shown in Figure 3, the net effect of increasing
the load current is reduced PSRR.
The differential DC voltage between input and output
also affects PSR. As V
IN
– V
OUT
is lowered, the pFET
(which provides gain) is driven out of the active (satura-
tion) region of operation and into the triode/linear region,
which causes the feedback loop to lose gain. Therefore,
the PSR of the regulator decreases as V
IN
approaches
V
OUT
. The lowest PSR, approaching 0 dB, occurs when the
device is in dropout (V
IN
≈
V
OUT
). In this situation, the RC
filter formed by the linear regulator’s pass-element R
DS
and output capacitor determines PSR.
Low noise
A linear regulator’s self-generated noise is sometimes
confused with its PSRR. However, noise is generated by
the transistors and resistors in the regulator’s internal
circuitry as well as by the external feedback resistors.
Transistors generate shot noise and flicker noise, both of
which are directly proportional to current flow. Flicker
noise is indirectly proportional to frequency and so is
higher at low frequencies. The resistive element of
MOSFETs also generates thermal noise like resistors.
Thermal noise is directly proportional to temperature, the
resistor’s resistance value, and the current flow through
the transistors. Transistors and resistors closest to the
error-amplifier inputs in the small-signal path cause the
Figure 3. TPS717xx PSRR graph
80
150 mA
70
10 mA
60
50
40
75 mA
30
20
V
IN
– V
OUT
= 1 V
C= 1 F
C
10
0
OUT
NR
= 10 nF
10
100
1 k
10 k
100 k
1 M
10 M
Frequency (Hz)
18
Analog Applications Journal
High-Performance Analog Products
www.ti.com/aaj
3Q 2007
Texas Instruments Incorporated
Power Management
most output noise because their noise is amplified by the
regulator’s closed-loop gain (A
CL
= V
OUT
/V
Bandgap
= 1/
β
=
1 + R1/R2). The noise contribution from components later
in the signal path is insignificant when compared to the
noise at the error-amplifier inputs. In fact, when modest-
sized feedback resistors are used, most of the regulator’s
noise comes from the amplified bandgap reference. As
shown in Figure 1, the simplest way to reduce the bandgap
noise is to use a low-pass filter (LPF) consisting of an
internal resistor, R
LPF
, and an external capacitor, C
LPF
.
At startup, this filter would slow down the output-voltage
rise without the aid of the “quickstart” transistor. When
the quickstart transistor is used, it shorts out the R
LPF
for
a short time at startup so the regulator output can rise
quickly. Larger noise capacitors such as C
LPF
in Figure 1
will reduce the output noise produced by the bandgap
until the regulator’s other noise sources begin to dominate.
Using a noise capacitor that is too large results in the
quickstart circuit timer expiring before the R
LPF
×
noise-capacitor pin is not available, adding a capacitor
across R1 reduces the noise by reducing the closed-loop
gain at high frequencies. However, this could potentially
slow down start-up time, since the capacitor would have
to be charged by the current in the resistor divider; adding
such a capacitor could also potentially make the feedback
loop unstable.
Component selection and board layout
Proper board layout and capacitor selection are critical to
maximizing PSR and minimizing noise. Low-ESR output
capacitors maximize PSR at high frequencies but may
increase noise. The reason for this is that the low imped-
ance created by the output capacitor and its ESR may
improve stability and PSR by removing peaking in the
control loop at frequencies near f
P(dom)
, but removing this
peaking would also provide higher gain for the internal
noise sources. To maximize PSR and minimize noise, it is
recommended that V
IN
and V
OUT
have separate ground
planes that are connected at the regulator’s ground pin.
The input, output, and noise-reduction capacitors should
be very close to the IC, with the ground of the noise-
reduction capacitor as close to the regulator’s ground pin
as possible.
Conclusion
Linear regulators are ideal for providing a low-ripple,
low-noise power rail to sensitive analog circuitry. The
TPS717xx single-output and TPS718xx/TPS719xx dual-
output linear regulators are specifically designed for
providing high PSR over a wide frequency range with low
noise. These linear regulators also consume very little
quiescent current when powered and even less when shut
down, helping maximize battery life in portable powered
applications that need bursts of regulator power only at
irregular intervals.
References
For more information related to this article, you can down-
load an Acrobat Reader file at www-s.ti.com/sc/techlit/
litnumber
and replace
“litnumber”
with the
TI Lit. #
for
the materials listed below.
C
LPF
time constant. In this case, the output voltage will rise
quickly to a level below regulation and then rise very
slowly to its final regulated value.
A regulator’s noise output is characterized by two mea-
surements. One is its spectral noise density, a curve that
shows noise (µV/
H
—–
) versus frequency. The other is the
RMS of the spectral noise density integrated over a finite
frequency range, also commonly called output-noise volt-
age (µV
rms
). Figure 4 shows the TPS717xx’s spectral noise
density with different C
NR
values, where C
NR
is the same
as C
LPF
in Figure 1.
When noise specifications of different regulators are
compared, it is imperative that the two regulators’ noise
measurements be taken over the same frequency range
and at the same output voltage and current values. When
output noise values for regulators at two different output
voltages are compared, an approximate noise value can be
used that is computed by scaling one of the noise measure-
ments by the ratio of the two output voltages. When a
√
Figure 4. TPS717xx spectral noise density
Document Title
TI Lit. #
30
1. Vishal Gupta, Gabriel A. Rincón-Mora, and
Prasun Raha, “Analysis and Design of
Monolithic, High PSR, Linear Regulators for
SoC Applications,”
http://users.ece.gatech.edu/
rincon-mora/publicat/journals/socc04_psr.pdf
I
= 10 mA
OUT
OUT
=1µF
25
C
20
—
15
2. John C. Teel, “Understanding noise in
linear regulators” . . . . . . . . . . . . . . . . . . . . . . . .
.slyt201
3. John C. Teel, “Understanding power supply
ripple rejection in linear regulators” . . . . . . . . .
.slyt202
Related Web site
power.ti.com
10
C=
10 nF
NR
C
=0nF
NR
C=
100 nF
C
=1nF
5
NR
NR
0
100
1k
10 k
100 k
Frequency (Hz)
19
Analog Applications Journal
3Q 2007
www.ti.com/aaj
High-Performance Analog Products
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